One of the nice things about the uTracer testimonial page is that I get a bit of an insight into the sort of tubes people want to trace. From the testimonials I received from people who built the uTracer6, I was astonished to see what “monster tubes” especially ham radio enthusiasts are using! Whereas I would have thought that the uTracer6 with its 1000 V and 1 A anode and screen supplies would cover all possible audio tubes, it appears that for most transmitter tubes the uTracer6 only covers a small part of their operation area. Not surprising, just like Dickens’ Oliver Twist, the proud owners of these tubes are asking “Please Sir, can I have some more!” I took it as a challenge to think about circuit concepts that will allow for full tracing of these monster tubes. It should be clear that, although I may do some experiments to test concepts, I have at this moment no plan whatsoever to make a uTracer kit for these very high scary voltages. Also because I do not want to be held liable should something go wrong!
Figure 1.1 Some pictures from my testimonial page of the monster transmitter tubes being traced with the uTracer6.
Although with its 1000 V anode voltage the uTracer6 makes a good start, most people like Oliver Twist “would like to have some more”!
I made a small search on internet into the type of tubes ham radio amateurs use in their transmitter amplifiers. A small collection of them can be found in Fig. 1.2. It appears that anode voltages up to 5 kV at a current between 1 and 2 A would cover most tubes.
Figure 1.2 I made a small search on the internet into the type of tubes ham radio amateurs use in their transmitter amplifiers. It appears that anode voltages up to 5 kV at a current between 1 and 2 A would cover most tubes.
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My first idea was to revive the transformer idea again that I explored - and for various reasons abandoned - in the uTracer4 concept. The main attraction of that concept is its robustness. It appeared to be almost impossible to damage the circuit by a shorted output. Another attractive feature that I had in mind was to make the concept modular, so that to achieve higher currents more units could be placed in parallel. I looked at it with my colleague and transformer specialist Peter Bancken. We both agreed that that the concept could work, however, it would require a specially made transformer that had to combine a high output voltage with a very low spreading inductance, requirements that are difficult to combine.
Figure 2.1 Principle of the Marx impulse generator (left); giant Marx generators (right) can produce pulses of millions of Volts and are used, for example, to test components in electricity distribution networks.
The idea of making something modular, where either higher currents or voltages can be obtained by switching units in parallel or in series somehow reminded me of the giant Marx impulse generator we had at the high voltage lab of my university in Eindhoven. From Wiki: a Marx generator is an electrical circuit first described by Erwin Otto Marx in 1924. Its purpose is to generate a high-voltage pulse from a low-voltage DC supply. Marx generators are used in high-energy physics experiments, as well as to simulate the effects of lightning on power-line gear and aviation equipment.
A Marx generates a high-voltage pulse by charging a number of capacitors in parallel, and then suddenly connecting them in series. Figure 2.1 shows the basic circuit. First, n capacitors C are charged in parallel to a voltage Vhv by a DC power supply through the resistors R (Fig. 2.1A). The spark gaps are used as switches, and have the voltage Vhv across them, but the gaps have a breakdown voltage greater than Vhv, so they all behave as open circuits while the capacitors charge. To create the output pulse, the first spark gap is caused to break down (Fig. 2.1B); the breakdown effectively shorts the gap, placing the first two capacitors in series, applying a voltage of about 2*Vhv across the second spark gap. Consequently, the second gap breaks down to add the third capacitor to the "stack", and the process continues to sequentially break down all the gaps (Fig. 2.1C). With all capacitors in series, an output pulse of n*Vhv is obtained at the last stage. The charge available is limited to the charge on the capacitors, so the output is a brief pulse as the capacitors discharge through the load. At some point, the spark gaps stop conducting, and the voltage supply begins charging the capacitors again.
Searching for “Marx generator” in Google will return many hits of both professional as well as amateur builders of Marx generators, obviously with the sole purpose to produce giant sparks. I especially like the YouTube video from ElectroBOOM!
A Marx uTracer?
Figure 2.2 Schematical representation of the modular high voltage uTracer.
Since the uTracer also operates in a pulsed fashion, it seemed obvious to explore if the Marx concept can be adopted for a very high voltage uTracer. One very attractive aspect of the Marx generator is that during its operation both the resistors as well as the capacitors are never subjected to a voltage higher than the input voltage. This is very attractive since above 1500 V electronic components become very scarce and prohibitively expensive. A major difference with the standard Marx generator is that in the uTracer we need to interrupt the current after a fixed period of time of 1 ms, while in the Marx generator the current continues to run until the capacitors are so far discharged that the spark gaps open again. After some puzzling I came to a circuit that might do the trick.
The basic operation of the circuit is schematically explained in Fig. 2.2. At the center of the circuit in Fig. 2.2A we find capacitors C1 – C4, which obviously form the heart of the Marx generator. Note that on the bottom side of the ladder structure the resistors have been replaced by switches S1. On the top side of the ladder structure the resistors have been replaced by diodes. On the left side the ladder structure is not directly grounded, but connected to ground by a small series resistor that obviously will be used to measure the current during the measurement pulse. The DC voltage source in the traditional Marx generator has been replaced by a boost converter consisting of T1, L1 and D0. On the right side of the ladder switch S4 has been added to discharge the capacitors through Rd when needed. Finally, the spark gaps have been replaced by switches S2.
(the letters refer to the corresponding figure in Fig. 2.2)
Towards a circuit
It will not come as a surprise that in the real circuit implementation that I have in mind the switches are replaced by transistors. The obvious challenge will be to find a simple way to control the transistors. Let’s first discuss voltages. To keep the total circuit as small as possible the voltage per ladder stage needs to be as high as possible. The SiC SCT2750NY transistors are rated at 1700 V, combined with an Rdson of 0.75 ohm and a continuous drain current of 6 A! For me they are the ideal candidate for all the high voltage transistors. To be on the safe side, the maximum voltage per ladder stage was set at 1500 V.
Figure 2.3 Simplified circuit diagram of the modular high voltage uTracer.
Figure 2.3 shows a simplified version of the actual circuit implementation. Going from left to right we first find the traditional booster configuration that is used in the other uTracers. To reach the 1500 V working voltage I plan to use three 330uF / 500V electrolytic capacitors in series. To ensure that the total voltage distributes evenly over the three capacitors, usually a resistive divider network is used. However, selecting the proper resistor values is tricky and besides this the resistors also dissipate. So here I plan to use Zener diodes that limit the voltage per capacitor to a value slightly above 500 V. When the voltage across a capacitor exceeds the maximum value the diode starts to conduct, forcing more current into the other caps.
Next, we move to the ladder sections. Transistors Ta obviously implement the switches S1 from Fig. 2.2. The gates of these transistors are controlled by a bus labelled “Open ladder.” This is an active-low signal. When it is 0 V, diodes Da block and the gates of the transistors are pulled low (with respect to their sources) by a resistor. When it rises to 20 V, first the most left transistor whose source is practically connected to ground via Rs will start to conduct. This will also pull the source of the second transistor Ta to ground so that also that transistor will turn on, etc. When the ladder is “erected,” and the different sections will be at a high potential, diodes Da will block, and the transistors Ta will be off. This means that some of the diodes Da will need to be able to block the maximum output voltage of the circuit! The high resistivity resistors Rl have been added to prevent the different branches of the ladder from “floating” when the transistors Ta are open.
When the ladder is in the “charging state,” the total equivalent high voltage capacitance for the circuit in Fig. 2.3 is: 3 x (330 uF / 3) = 330 uF. The first question now is: is it possible by using a simple boost converter, to charge a 330 uF capacitor to 1500 V in a reasonable time? I will come back to that question later. Apart from this, in the “erected state” all the capacitors are in series, and we end up with an equivalent capacitance of 330 uF / 9 = 37 uF. This is not so far off from the 50 uF in the uTracer6, but it is getting a bit on the small side. Up till now I have used the 10 ADC inputs of the microcontroller to sequentially poll the analog inputs. However, this inevitable means that there will be a time difference between the moment the voltage is measured and the moment the current is measured. In between these moments, the voltage of the capacitors may have dropped significantly. So for the first prototype I plan to use sample-and-holds, that capture both the voltage as well as the current at exactly the same moment.
The transistors Tb in Fig. 2.3 implement the switches S2 in Fig. 2.2. People who have studied the circuit diagram of the uTracer6 will undoubtably recognize the gate driver circuit for Tb. The circuit makes use of the fact that “bottom side” of the ladder will be at ground potential most of the time. In this state capacitor Cb will be charged to 20 V through diode Db. The capacitor acts as a little battery to power the gate driver circuit during the 1 ms measurement pulse. When transistors Tb are not conducting, their gates are tied to their sources by a resistor. The transistors are controlled by optocouplers that uses the charge stored in capacitors Cb. For the optocouplers I plan to use the affordable CNY66 that isolates voltages up to 13.5 kV.
On the right side of Fig. 2.3 we find the discharge circuit and the final output switch. In the discharge circuit transistor T2 fulfills two tasks. First, T2 is used to discharge capacitors Ca (if needed) through resistor Rd. At the same time T2 grounds capacitor Cb so that it can be charged. The output switch is identical to the ladder switches.
Figure 2.4 Detailed circuit diagram of the modular high voltage uTracer.
The open spaces in Fig. 2.3 already suggest that this is not the complete circuit. Correct! The way the gates of the transistors are driven in Fig. 2.3 is a bit too simple. The grid resistor that pulls the gate down unavoidably has a rather high impedance. Any fast transient on the drain can, through the drain-gate capacitance, pull the transistor into conduction with potentially disastrous consequences (read more about the so called dV/dt breakdown here). To tie the gates of the transistors firmly to ground when they should remain closed, totem pole buffers consisting of an emitter follower npn/pnp pair have been added. Actually, this small circuit is identical to the high-voltage switch driver already tested in the uTracer6.
Figure 2.5 Ta and Tb are protected by the Zener diodes across the electrolytic capacitors.
Although in principle the high voltage transistors in the circuit are never subjected to a voltage higher than the maximum voltage of a single ladder stage, it turned out that the Zener diodes used to guarantee the proper voltage distribution over the electrolytic capacitors also offer additional protection. When for example the voltage across Tb in Fig. 2.5a would become too high, the voltage is clamped by the stack of Zener diodes on the right side of the stage. Conversely, when the voltage across Ta becomes too high (Fig. 2.5b), the “body diode” of Tb opens and the voltage over Ta is clamped by the stack of Zener diodes over the capacitors on the left side of the circuit. The body diode is in circuit diagrams often omitted because in most circuits the drain always remains positive with respect to the source, however, here it is used to offer extra protection to the circuit. In the case of the SCT2750N the characteristics of the body diode are separately specified in the datasheet. It can handle a continuous current of 6 A and a peak current of 14 A.
Figure 2.6 Boost converter variations
It can easily be overlooked that also the inductor in the boost converter must be able to handle the full ladder stage voltage of 1500 V. From the extensive experience with the uTracer3 and 6 it is clear that the inductors from the DR127 series from Coiltronics / Eaton can perfectly handle voltages up to 500 V. By simply placing a number of inductors in series the maximum voltage can be distributed over the separate inductors. Since the induction voltage over the inductors is directly proportional to dI/dt, and since the same current flows through all the inductors, the voltage will always be evenly distributed. In the uTracer6 two inductors were used in series to be able to handle 1000 V. In this case three inductors of 100 uH can be used to roughly obtain the 330 uH to match the pulse length of the firmware controlled boost converters.
Mainly for my own administration a list of components I have in mind for the 5 kV uTracer:
Figure 2.7 A first experiment to test the charging of a battery of 330uF / 500 V capacitors ended in a big bang! The capacitors were switched as three banks of three capacitors in series adding up to a voltage of 1500 V (@ 110 uF). The idea to use two UF4007 diodes in series in the boost converter (see Fig. 2.6) was apparently not a good idea as one of the diodes exploded with an enormous bang that made my ears whistle for the better part of an hour!
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While experimenting with the circuit of the 5 kV uTracer discussed in the previous section, something went wrong which left the high voltage reservoir capacitors partly charged. Fortunately, I thought of it, could discharge them safely, but had I forgotten to check them, I could have been in for a very nasty surprise! Not wanting to rely on my alertness to continuously check the status of the capacitors during experiments, I felt the need for a circuit that will warn me if the capacitors are charged to a dangerous voltage. It resulted in a simple circuit that might be of interest to other uTracer owners who want to add some safety to their uTracer builds.
Below the design requirements for the lifesaver:
Figure 3.1 The conception of the lifesaver step-by-step.
Obviously a circuit that makes a current of a few micro-amps flash an LED will need to consist of a capacitor and some form of threshold detector with switch that discharges the capacitor through the LED. The basic idea of the circuit is shown in Fig. 3.1A. It consists of two parts, a current source that limits the current drawn by the circuit to a few micro-amps and makes it independent of the input voltage, and the flasher circuit.
Let’s first concentrate on the flasher circuit. The simple idea was to let the current charge capacitor C1 until the voltage over the capacitor becomes so high that zenerdiode D1 starts to conduct thereby triggering thyristor D2. The thyristor discharges the capacitor through the LED until the capacitor is discharged, and the thyristor comes out of conduction. The low power thyristor can conveniently be implemented by a pnp transistor tightly coupled to an npn transistor Fig. 3.1B (also have a look here). The problem with this circuit is that the gain of common transistors is so high that the smallest noise in the circuit will trigger the thyristor at unpredictable moments. The problem is easily solved by connecting two high value resistors over the emitter-base junctions of the two transistors Fig. 3.1C. This will effectively kill the gain of the transistors at low currents, making it a reliable simulated thyristor.
Disappointingly, the circuit still did’t work as expected. The capacitor charged as it should, and when the capacitor voltage increased beyond the voltage of the zener diode the thyristor fired as expected flashing the LED and at the same time discharging C1. However, instead of starting the cycle all over again, the circuit froze at the point where the current supplied to the circuit would just keep the thyristor conducting. The solution to this was to include a small inductor in series with the LED (see Fig. 3.1D). The inductor in combination with the capacitor forms a resonant circuit that gives the circuit a flywheel character whereby the current is really pushed to zero so that thyristor comes out of conduction.
The circuit of Fig. 3.1D worked well, but only for reasonable currents. For currents in the few micro-amp range the circuit again froze in a state where the current needed to trigger the thyristor was larger than the current supplied to the circuit. So for this circuit to work, the current drawn during the charging of the capacitor should be an order of magnitude or more less than the supply current. This was easily remedied by the addition of MOSFET T3 (Fig. 3.1E). With zenerdiode D1 in series with the gate, T3 will only start to conduct when the gate voltage exceeds its threshold voltage plus the zenerdiode voltage, while the gate of T3 of course draws zero current. Note that in this case the thyristor is not trigger by pulling the base of T2 up, but by pulling the base of T1 down. This really did the trick and the circuit now works very satisfactory even for currents well below 1 uA.
Before attentive readers start writing me emails, I know that there are special energy scavenging ICs like the LTC3588 or the S6AE101A, but that would have been half so much fun, and they are quite expensive anyway.
Figure 3.2 Left, connection of the lifesaver circuit to the uTracer in case each capacitor has its own lifesaver circuit. Right, connection if one circuit is used to monitor both capacitors.
What remains to be discussed is the current source. For that I used a circuit that most of you will recognize as the current limiting circuit in many simple power supplies. It consists of transistor Ta that is basically pulled into conduction by resistor Ra (topside of Fig. 3.1B). Rb senses the current and as soon as the current exceeds approximately 0.8 V, transistor Tb will start to conduct thereby cutting off Ta. This will stabilize the current at a value determined by Rb. I used a MOSFET for Ta since I didn’t want to bother with a base current and I had some lying around anyway. Diode Da protects the gate of the MOSFET against an excessive gate voltage should anything go wrong.
In Fig. 3.2 some component values have been added to the circuit. For the 1000 V version of the circuit an FQD2N100 or STD2NK100 or equivalent can be used for Ta. If you want to use the lifesaver for the uTracer3 obviously a transistor with a lower breakdown voltage can be selected. Resistor Ra should be as high as possible since the current through this resistor adds to the total current making the current source voltage dependent. I used a 500 Mohm resistor from Murata (available from Mouser). The MHR_SA series of resistors from Murata offers resistors up to 1 Gohm with up to 20 kV rating for a very reasonable price. For the inductor I used a 100 uH axial type, the value is not particularly critical. I don’t know where I got this particular inductor from, but I guess that any readily available inductor like this one available from Mouser will do. For the LED a high intensity type gives the best results. I used a 3500 mcd white VAOL-3GWY4 from VCC.
Figure 3.2 shows how the lifesaver can be connected to the uTracer. If you plan to use a seperate lifesaver circuit for each of the two high voltage electrolytic capacitors the circuit can be connected directly over each of the capacitors (Fig. 3.2A). In this way, the current used by the circuit is not flowing through the current sense resistor and so is not detected at all. It is also possible to use one circuit for both capacitors by using two diodes in an “or” configuration (Fig. 3.2B). Obviously, in that case it is unknown which of the two capacitors (or both) is charged.
The (rather poor) movie below shows the circuit in action on breadboard. I am not going to bother designing a PCB for it, but will simply transfer it to perfboard(s). It is fun to experiment with the circuit. Increasing the value of C1 will increase the intensity of the flash but reduce the flash frequency. This can be compensated for by increasing the current by lowering the value of Rb. The voltage at which the circuit starts flashing can be adjusted by changing the breakdown voltage of the zener diode, which will also effect the brightness of the flash etc etc. In the test setup shown in the movie below I connected a piezo speaker in parallel with the LED to also give an acoustic feedback signal. Perhaps a small normal speaker in series with the LED will also work and provide enough inductance so that L1 is not needed? In short, have fun but be careful with high voltages!
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The first experiments with the Marx generator inspired idea for a very high voltage uTracer turned out to be rather a disappointment. The first experiments to see if it is feasible to charge a battery of electrolytic capacitors with a simple boost converter ended literally with a big bang when a UF4007 diode in one of the boost converter branches exploded. Although I still think there is some merit in the concept, I came to the following thoughts / conclusions.
First, very high voltage capacitors with any significant capacitance are very scarce. In practice 500 V rated electrolytic capacitors are at the upper range of what is commercially available. So, to get to a higher voltage, stacking (placing in series) of capacitors is the only option. This is not straightforward. Special measures are needed to ensure that capacitors in series are charged equally, even in case they have small differences in leakage currents. On top of that the total capacitance of the stack decreases inversely proportional to the number of capacitors stacked.
Secondly, if we want to go to higher voltages and currents, it becomes almost unavoidable to go to shorter measurement pulses. In the first place just to reduce the volume (and cost) of the storage capacitors, but in the second place also from a safety point of view. In the capacitor bank shown in Fig. 2.7 I had 9 capacitors of 330 uF all charged to 500 V. Such a monster can kill you if you are unlucky! Going to shorter measurement pulses automatically means that we cannot simply successively readout the analog signals with the single AD converter of the PIC anymore, but that we will need sample-and-holds, perhaps even in combination with multiple AD converters to capture voltages and currents accurately in the same moment of time.
Turning things around
Figure 4.1 Instead of applying a voltage and measuring the resultant current (A), it is just as well possible to force a current and measure the resultant voltage (B).
My normal work, and the demand for uTracer kits (and the associated desperate searching for the necessary components) kept me pretty busy the past months. Nevertheless, the thoughts about the 5 kV / 1 A version are never far away, especially not at night. At one of these moments it occurred to me that it might perhaps be advantageous to turn things around, and not use a capacitor to store the energy for the measurement pulse, but an inductor! A capacitor stores energy in the form of charge that at the terminals is available as a voltage that, connected to a load, will result in a current. An inductor stores energy in the form of magnetization that at the terminals is available as a current that, connected to a load, will result in a voltage (Fig. 4.1).
Figure 4.2 Left (A1&A2), basic working of the uTracer3 & 6. Right (B1&B2) principal idea of the “turned around” concept.
This arrangement has several advantages:
Already from the onset a few disadvantages of the new circuit are clear, and I am pretty sure that going along I will discover a few more:
A first shot
Somehow it occurred to me that the inductive ballasts that were (are) used in fluorescent lamps might be a good candidate inductor for the first experiments. They have a significant inductance in the order of 1 H and are designed to sustain a voltage well in excess of 1000 V. Unfortunately, I didn’t have any lying around, but fortunately my colleague Peter Blanken just replaced all the fluorescent lights in his house with LED lights and he was so kind to donate four of them to the cause of tube tester research.
It is very simple to determine the inductance of this type of large inductors. First, simply measure the DC resistance Rdc. Then connect the inductor to an AC voltage. I my case I used a 20 V transformer. Measure the voltage across the inductor terminals and the current flowing through the inductor. This yields the total impedance of the inductor Z = V/I. The inductance now follows from L = (Z-Rdc)/(2*pi*freq). The inductors I had were all designed for 36-40 W / 230 V fluorescent lamps, and despite the fact that all four of them came from different manufacturers, they all had a DC resistance close to 43 ohm, and an inductance around 1.3 H. Apart from the DC resistance and the inductance the saturation current is an important parameter. More about that in the next section.
Figure 4.3 Circuit diagram of the test circuit.
Figure 4.4 Breadboard setup.
Just to get a first hands-on feeling on how the circuit with such a ballast inductor would function, I breadboarded a quick and dirty test setup (Fig. 4.3 and 4.4). The circuit consists of a simple debounced pushbutton, and a one-shot. The pulse width of the one-shot can be crudely set by changing capacitor C1. The pulse from the one-shot drives the gate of a SCT2750 SiC MESFET. This SiC monster transistor combines a breakdown voltage of 1700 V with an on-resistance of 0.75 ohm. It is the same transistor that is used in the boost converter of the uTracer6. The circuit around the inductor is powered by a floating, well decoupled power supply. The power supply is connected to ground via a small current sense resistor that is used to monitor the current through the inductor. Note that in this arrangement the current will cause a negative voltage drop over the resistor with respect to ground!
Capacitor C3 is added to broaden the width of the high-voltage pulse. When the inductor is “charged,” and T1 is opened part of the current will charge C3. When the voltage has reached its maximum the current through C3 will reverse, and it will start dumping its charge into the load. Of course, this somewhat reduces the output voltage, but it makes life for the sample-and-hold circuits much easier. There is another benefit! With a small series resistance in the ground connection of C3 the current reversal at the maximum output voltage can be detected. This can be used to trigger the sample-and-hold circuits. Zener diode D2 is necessary to ensure that during the off-state, and the inductor charging phase the current though the load is zero. The breakdown voltage of the Zener diode must be higher than the power supply voltage Vb. In this experiment I had a 120V / 1W diode lying around that I used. For the load I used a heavy duty 1k / 5W power resistor. At 1 A current the output voltage will be 1000 V what I consider to be on the safe side for the inductor.
Figure 4.5 “Charging” the inductor with three different voltages. The current is measured by measuring the voltage drop over resistor Rs. Because this voltage drop is measured with respect to ground it has a negative sign. The small triangle indicates the trigger point which coincides with the moment T1 is opened. In the top row measurements the time base it set to 20 ms to register the charging process. In the bottom row measurements the time base is set to 200 us to zoom in to the discharging of the inductor.
The first thing to have a look at is the saturation behavior of the inductor. The simplest way to do that is to apply a constant voltage to the inductor and monitor the current. For a perfect inductor the current will increase linear with time (∆I = (V/L)* ∆t) until the point of saturation. At that point all the magnetic domains in the iron core of the inductor will be completely aligned with the magnetic field so that they cannot contribute to the inductance anymore with a further current increase. Basically, from that point onward, the core of the inductor is out of the equation, and what remains is an air inductor with a much smaller inductance resulting in a sharp increase in current.
In our inductor the situation is a bit more complex because of the relatively high series resistance of the inductor, which tends to obscure the saturation behavior. The snapshots in Fig. 2.5 show the current through the inductor as a function of time as measured over series resistance Rs. Because of the direction of the current this voltage drop is negative with respect to ground. With a maximum voltage of my power supply of 50 V, it takes about 70 ms to charge the inductor to approximately 1 A. The three columns Fig. 2.5 show measurements at supply voltages 32 V, 42 V and 50 V respectively. The two rows represent the same measurements, but at a different timescale. Note that none of the top row measurements shows the sharp increase in current we would expect in case of saturation. For column A I am pretty sure the core has not saturated yet. In Fig. B1 we clearly see the current leveling off. I am convinced the core starts to saturate at this point. From that point onwards, the current remains constant, and is completely determined by the supply voltage and the series resistance. Note that Vb / Rs = 42 / 44 = 0.95 A ≈ 0.930 A. More telling are the zoomed-in Figs. B2 and C2. The curves left to the trigger point (center) appear to be constant, because we are now looking at a 100x smaller time scale. To the right of the center we see the discharging phase of the inductor. Note that the first part of this discharge in Figs. B2 and C2 progresses much faster (encircled) than in remaining part of the curve. I assume that at that moment the core is still saturated, and we only see the small inductance of the air coil resulting in a much smaller time constant. At a certain point, basically corresponding to the current level of Fig. A2, the inductor comes out of saturation and the discharging continues at a much slower pace.
In short, our ballast inductor can certainly be used to a current of ≈ 700 mA, but as we will see in the following even 1 A is certainly possible.
Figure 4.6 Voltage at the drain of T1 for three different values of C3.
Figure 4.7 LTspice simulation of the basic circuit. Click here to download the LTspice input file.
Figure 4.7 shows an LTspice simulation of the circuit. In the simulation T1 is replaced by an ideal switch. Apart from the series resistance of the inductor no losses have been considered, and also the saturation of the inductor is not modeled. Despite this, the simulated version of the circuit shows a good agreement with the measurements. The simulations revealed quite pronounced oscillations that occur after the voltage has dropped to the level when the Zener diode stops conducting. Also, the measurements show the oscillations with a measured frequency of approximately 463 Hz, which corresponds perfectly with the resonance circuit formed by inductor L and C3 (L1 and C2 in the simulation).
Conclusions and outlook
Time for some conclusions and some ideas on if and how to continue:
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